ETSI TR 102 443 V1.1.1 (2008-08)

Technical Report
Satellite Earth Stations and Systems (SES);
Satellite Component of UMTS/IMT-2000;
Evaluation of the OFDM as a Satellite Radio Interface

2 ETSI TR 102 443 V1.1.1 (2008-08)

Reference
DTR/SES-00252
Keywords
satellite, UMTS
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ETSI
3 ETSI TR 102 443 V1.1.1 (2008-08)
Contents
Intellectual Property Rights.5
Foreword.5
1 Scope.6
2 References.6
2.1 Normative references.6
2.2 Informative references.6
3 Definitions, symbols and abbreviations .7
3.1 Definitions.7
3.2 Symbols.8
3.3 Abbreviations.8
4 OFDM technology and background.9
4.1 OFDM Fundamentals.9
4.1.1 OFDM Definitions.9
4.1.2 OFDM Signal Generation.10
4.1.3 Guard Interval.11
4.1.4 Impact of Guard Interval.12
4.1.5 Impact of Symbol Duration .12
4.1.6 Impact of Inter-Carrier Spacing .12
4.1.7 OFDM Inactive Sub-Carriers.12
4.1.8 Time-Frequency Multiplexing.13
4.1.9 OFDM Signal Reception Using the FFT .14
4.2 OFDM for Mobile Terrestrial and Satellite Scenario .14
5 OFDM and the satellite environment .15
5.1 Non-Linearity Effects and Predistortion Techniques .15
5.1.1 Compensation Techniques.15
5.1.2 Digital Predistortion Techniques .16
5.1.3 Multi-Beam Coverage Using OFDM.16
6 OFDM feasibility .17
6.1 Physical Layer Structure in the OFDM Downlink .17
6.1.1 Physical Channel.17
6.1.1.1 OFDM Physical Channel Definition .18
6.1.2 Channel Coding and Multiplexing.19
6.1.3 Physical Channel Mapping.20
6.1.4 User Traffic Multiplexing Solutions.20
6.1.4.1 Solution based on a generic Costas sequence .20
6.2 Spectrum Compatibility.22
7 OFDM Evaluation Scenario .23
7.1 Reference System Scenario for OFDM S-DMB Analysis.23
7.2 Reference OFDM configurations for the evaluation .24
8 Simulation Results.25
8.1 Uncoded System Performance .25
8.1.1 AWGN Channel.25
8.1.2 Non-linear channel.26
8.2 WCDMA Coding Performance .27
8.2.1 Non selective Rice fading .29
8.2.2 Frequency Selective Channel.30
9 Link Budget Study .36
9.1 System parameters.36
9.1.1 Satellite parameters.36
9.1.2 UE parameters.36
9.1.3 Physical layer configuration and performances .36
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4 ETSI TR 102 443 V1.1.1 (2008-08)
9.2 Link budgets.37
9.2.1 Handset.37
9.2.2 Handheld.38
9.2.3 Vehicular.39
10 Conclusions.39
History .41

ETSI
5 ETSI TR 102 443 V1.1.1 (2008-08)
Intellectual Property Rights
IPRs essential or potentially essential to the present document may have been declared to ETSI. The information
pertaining to these essential IPRs, if any, is publicly available for ETSI members and non-members, and can be found
in ETSI SR 000 314: "Intellectual Property Rights (IPRs); Essential, or potentially Essential, IPRs notified to ETSI in
respect of ETSI standards", which is available from the ETSI Secretariat. Latest updates are available on the ETSI Web
server (http://webapp.etsi.org/IPR/home.asp).
Pursuant to the ETSI IPR Policy, no investigation, including IPR searches, has been carried out by ETSI. No guarantee
can be given as to the existence of other IPRs not referenced in ETSI SR 000 314 (or the updates on the ETSI Web
server) which are, or may be, or may become, essential to the present document.
Foreword
This Technical Report (TR) has been produced by ETSI Technical Committee Satellite Earth Stations and Systems
(SES).
ETSI
6 ETSI TR 102 443 V1.1.1 (2008-08)
1 Scope
The present document entails a feasibility study that evaluates the use of the OFDM Radio Interface proposed the
3GPP TR 25.892 [i.1] as Satellite Radio Interface on the satellite downlink, presenting physical layer results and link
budget studies. The present document contains informative elements that should serve as a starting point for the
definition and finalization of advanced Satellite Radio Interfaces. The adoption of the OFDM Radio Interface results in
higher link margin under key propagation conditions such as the NLOS propagation case and when CGCs are
considered.
2 References
References are either specific (identified by date of publication and/or edition number or version number) or
non-specific.
• For a specific reference, subsequent revisions do not apply.
• Non-specific reference may be made only to a complete document or a part thereof and only in the following
cases:
- if it is accepted that it will be possible to use all future changes of the referenced document for the
purposes of the referring document;
- for informative references.
Referenced documents which are not found to be publicly available in the expected location might be found at
http://docbox.etsi.org/Reference.
For online referenced documents, information sufficient to identify and locate the source shall be provided. Preferably,
the primary source of the referenced document should be cited, in order to ensure traceability. Furthermore, the
reference should, as far as possible, remain valid for the expected life of the document. The reference shall include the
method of access to the referenced document and the full network address, with the same punctuation and use of upper
case and lower case letters.
NOTE: While any hyperlinks included in this clause were valid at the time of publication ETSI cannot guarantee
their long term validity.
2.1 Normative references
The following referenced documents are indispensable for the application of the present document. For dated
references, only the edition cited applies. For non-specific references, the latest edition of the referenced document
(including any amendments) applies.
Not applicable.
2.2 Informative references
The following referenced documents are not essential to the use of the present document but they assist the user with
regard to a particular subject area. For non-specific references, the latest version of the referenced document (including
any amendments) applies.
[i.1] 3GPP TR 25.892 (V6.0.0): "3rd Generation Partnership Project; Technical Specification Group
Radio Access Network; Feasibility Study for Orthogonal Frequency Division Multiplexing
(OFDM) for UTRAN enhancement (Release 6)".
[i.2] 3GPP TR 25.858 (V5.0.0): "3rd Generation Partnership Project; Technical Specification Group
Radio Access Network; High Speed Downlink Packet Access: Physical Layer Aspects
(Release 5)".
ETSI
7 ETSI TR 102 443 V1.1.1 (2008-08)
[i.3] ETSI TS 125 212: "Universal Mobile Telecommunications System (UMTS); Multiplexing and
channel coding (FDD) (3GPP TS 25.212 version 5.9.0 Release 5)".
[i.4] S. Chang: "Compensation of nonlinear distortion in RF power amplifiers", Wiley Encyclopedia of
Telecommunications, J.J. Proakis Ed., 2002.
[i.5] S. Benedetto and E. Biglieri: "Nonlinear equalization of digital satellite channels", IEEE J. Select.
Areas Comm., vol. 1, pp. 57-62, Jan. 1983.
[i.6] J.K. Cavers: "Amplifier Linearization using a digital predistorter with fast adaptation and low
memory requirements", IEEE Trans. Vehic. Tech., vol. 39, pp. 31-40, Nov. 1990.
[i.7] P. Salmi, M. Neri, and G.E. Corazza: "Fractional Predistortion. Techniques with Robust
Modulation Schemes for Fixed and mobile Broadcasting", 13th IST Mobile & Wireless
Communications Summit (IST2004), pp. 990-995, June 2004.
[i.8] S.W. Golomb, and H. Taylor: "Construction and Properties of Costas Array", Proc. IEEE, vol. 72,
pp. 1143-1163, Sep. 1984.
[i.9] S. Cioni, G.E. Corazza, M. Neri, and A. Vanelli-Coralli: "On the Use of OFDM Radio Interface
for Satellite Digital Multimedia Broadcasting Systems", International Journal of Satellite
Communications and Networking, February 2006, Int. J. Satell. Commun. Network. 2006; 24:153-
167, published online in Wiley InterScience (www.interscience.wiley.com). DOI: 10.1002/sat.836.
3 Definitions, symbols and abbreviations
3.1 Definitions
For the purposes of the present document, the following terms and definitions apply:
cell: geographical area under Complementary Ground Component coverage
downlink: unidirectional radio link for the transmission of signals from a satellite to a UE
forward link: unidirectional radio link for the transmission of signals from a gateway to a UE via a satellite
guard interval / guard time: number of samples inserted between useful OFDM symbols, in order to combat
inter-OFDM-symbol-interference induced by channel dispersion and to assist receiver synchronization
NOTE: It may also be used to aid spectral shaping. The guard interval may be divided into a prefix (inserted at
the beginning of the useful OFDM symbol) and a postfix (inserted at the end of the previous OFDM
symbol).
inter-carrier frequency / sub-carrier separation: frequency separation between OFDM sub-carriers, defined as the
OFDM sampling frequency divided by the FFT size
OFDM unit: group of constellation symbols to be mapped onto a sub-band, a subset of the OFDM carriers
OFDM samples: discrete-time complex values generated at the output of the IFFT, which may be complemented by
the insertion of additional complex values (such as samples for pre/post fix and time windowing)
NOTE: Additional digital signal processing (such as filtering) may be applied to the resulting samples, prior to
being fed to a digital-to-analog converter.
OFDM sampling frequency: total number of samples, including guard interval samples, transmitted during one
OFDM symbol interval, divided by the symbol period
repeater: device (e.g. CGC) that receives, amplifies and transmits the radiated or conducted RF carrier both in the
down-link direction (from the satellite to the mobile area) and in the up-link direction (from the mobile to the satellite)
return link: unidirectional radio link for the transmission of signals from a UE to a gateway via a satellite
rice factor: power ratio between LOS component and diffuse component
ETSI
8 ETSI TR 102 443 V1.1.1 (2008-08)
spot: geographical are under beam coverage
uplink: unidirectional radio link for the transmission of signals from a UE to a satellite
useful OFDM symbol: time domain signal corresponding to the IFFT/FFT window, excluding the guard time
useful OFDM symbol duration: time duration of the useful OFDM symbol
3.2 Symbols
For the purposes of the present document, the following symbols apply:
F OFDM sampling frequency
F Maximum Doppler shift.
d
N Total number of IFFT/FFT bins (sub-carriers)
N Number of prefix samples
p
N Number of modulated sub-carriers (i.e. sub-carriers carrying information)
u
T OFDM symbol period
s
T OFDM prefix duration
g
T OFDM useful symbol duration
u
Δf Sub-carrier separation
3.3 Abbreviations
For the purposes of the present document, the following abbreviations apply:
ACI Adjacent Channel Interference
APSK Amplitude and Phase Shift Keying
AWGN Additive White Gaussian Noise
BER Bit Error Rate
C/N Carrier to Noise power ratio
CGC Complementary Ground Component
CRC Cyclic Redundancy Check
CPICH Common Pilot Channel
DC-RF Direct Current to Radio Frequency
DL Down Link
EIRP Effective Isotropic Radiated Power
FDM Frequency Division Multiplexing
FFS For Further Study
FFT Fast Fourier Transform
FIR Finite Impulse Response
GEO Geostationary Earth Orbit
GW GateWay
HARQ Hybrid Automatic Repeat reQuest
HPA High Power Amplifiers
HSDPA High Speed Downlink Packet Access
HS-DSCH High Speed - Downlink Shared CHannel
IBO Input Back-Off
IFFT Inverse Fast Fourier Transform
IMR Intermediate Module Repeater
ISI Inter Symbol Interference
LOS Line-Of-Sight
LTWTA Linearized Travelling Wave Tube Amplifier
LUT Look-Up Table
MAC Medium Access Control
MIMO Multiple Input Multiple Output
NL Non Linear
NLOS No Line-Of-Sight
OBO Output Back Off
ETSI
9 ETSI TR 102 443 V1.1.1 (2008-08)
OFDM Orthogonal Frequency Division Multiplexing
PAPR Peak-to-Average Power Ratio
PDSCH Physical Downlink Shared CHannel
PER Packet Error Rate
PhCh Physical ChannelPSK Phase Shift Keying
QAM Quadrature Amplitude Modulation
SCCH Shared Control CHannel
S-DMB Satellite-Digital Mobile Broadcasting
SFN Single Frequency Network
SNR Signal-to-Noise Ratio
T-F Time-Frequency
TPCCH Transmit Power Control CHannel
TTI Transmission Time Interval
TWTA Travelling Wave Tube Amplifier
UE User Equipment
UTRAN UMTS Terrestrial Radio Access Network
WCDMA Wideband Code Division Multiple Access
4 OFDM technology and background
4.1 OFDM Fundamentals
4.1.1 OFDM Definitions
The technique of Orthogonal Frequency Division Multiplexing (OFDM) is based on the well-known technique of
Frequency Division Multiplexing (FDM). In FDM different streams of information are mapped onto separate parallel
frequency channels. Each FDM channel is separated from the others by a frequency guard band to reduce interference
between adjacent channels.
The OFDM technique differs from traditional FDM in the following interrelated ways:
1) multiple carrier multiple carriers (called sub-carriers) carry the information stream;
2) the sub-carriers are orthogonal to each other; and
3) a guard time may be added to each symbol to combat the channel delay spread and inter-symbol interference
induced by linear distortion.
These concepts are illustrated in the time-frequency representation of OFDM presented in figure 1.
5 MHz Bandwidth
FFT
Sub-carriers
Guard Intervals
…
Symbols
Frequency
…
Time
Figure 1: Frequency-Time representation of an OFDM Signal
Since the orthogonality is guaranteed between overlapping sub-carriers and between consecutive OFDM symbols in the
presence of time/frequency dispersive channels the data symbol density in the time-frequency plane can be maximized.
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10 ETSI TR 102 443 V1.1.1 (2008-08)
4.1.2 OFDM Signal Generation
Data symbols are synchronously and independently transmitted over a high number of closely spaced orthogonal
sub-carriers using linear modulation (either PSK, APSK or QAM). The generation of the QAM/OFDM signal can be
th
conceptually illustrated as in figure 2, where ω is the n sub-carrier frequency (in rad/s) and 1/T is the QAM symbol
n u
rate. Note that the sub-carriers frequencies are equally spaced and hence the sub-carrier separation is constant. That is:
ω − ω
n n−1
= Δf , n ∈[1, N −1] .
2π
In practice, the OFDM signal can be generated using IFFT digital signal processing. The baseband representation of the
th
OFDM signal generation using an N-point IFFT is illustrated in figure 3, where a(mN+n) refers to the n sub-channel
modulated data symbol, during the time period mT < t ≤ (m+1)T .
u u
QAM
modulator
.
j ω t
.
e
.
QAM
s ( t )
Σ
modulator
jω t
n
.
e
.
.
QAM
modulator
Symbol rate = 1/T ω
u j t
N −1
e
s
symbols/sec
Figure 2: Conceptual representation of OFDM symbol generation

mT
u
( m+1 )T
time u
a( mN + 0 )
mT
u
( m+1 )T
u
time
a( mN + 1 )
a( mN + 2 )
frequency
s (0), s (1), s (2), …, s ( N-1 )
m m m m
IFFT
.
.
s
. m
a( mN + N -1 )
Figure 3: OFDM useful symbol generation using an IFFT
The vector s is defined as the useful OFDM symbol. Note that the vector s is in fact the time superposition of the N
m m
narrowband modulated sub-carriers.
ETSI
11 ETSI TR 102 443 V1.1.1 (2008-08)
It is therefore easy to realize that, from a parallel stream of N sources of data, each one modulated with QAM useful
symbol period T , a waveform composed of N orthogonal sub-carriers is obtained, with each narrowband sub-carrier
u
having the shape of a frequency sinc function. Figure 4 illustrates the mapping from a serial stream of QAM symbols to
N parallel streams, used as frequency domain bins for the IFFT. The N-point time domain blocks obtained from the
IFFT are then serialized to create a time domain signal.
QAM symbol rate =
N/T symbols/sec
u
N OFDM
symbol
symbols Useful OFDM
QAM streams
N :1
IFFT
Source(s)
1: N
T symbols
1/ u
Modulator 1/T
u
symbols/s
symbol/sec
Figure 4: OFDM signal generation chain
4.1.3 Guard Interval
A guard interval may be added prior to each useful OFDM symbol. This guard time is introduced to minimize the
inter-OFDM-symbol-interference power caused by time-dispersive channels. The guard interval duration T (which
g
corresponds to N prefix samples) needs to be sufficient to cover the most of the delay-spread energy of a radio channel
p
impulse response. In addition, such a guard time interval can be used to allow soft-handover.
OFDM symbols
m
Prefix length
Useful OFDM symbol duration
copy
Figure 5: Cyclic prefix insertion
A prefix is generated using the last block of N samples from the useful OFDM symbol. The prefix insertion operation
p
is illustrated in figure 5. Note that since the prefix is a cyclic extension to the OFDM symbol, it is often termed cyclic
prefix. Similarly, a cyclic postfix could be appended to the OFDM symbol.
After the insertion of the guard interval the OFDM symbol duration becomes T = T + T .
s g u
The OFDM sampling frequency F can therefore be expressed as:
N + N
p
F =
T
s
hence, the sub-carrier separation becomes:
F
Δf = .
N
It is also worth noting that time-windowing and/or filtering is necessary to reduce the transmitted out-of-band power
produced by the ramp-down and ramp-up at the OFDM symbol boundaries in order to meet the spectral mask
requirements.
ETSI
12 ETSI TR 102 443 V1.1.1 (2008-08)
4.1.4 Impact of Guard Interval
The cyclic prefix should absorb most of the signal energy dispersed by the multi-path channel. The entire the
inter-OFDM-symbol-interference energy is contained within the prefix if the prefix length is greater than that of the
channel total delay spread, i.e.:
T >τ
g
where τ is the channel total delay spread. In general, it is sufficient to have most of the energy spread absorbed by the
guard interval, given the inherent robustness of large OFDM symbols to time dispersion, as detailed in the next clause.
4.1.5 Impact of Symbol Duration
The mapping of the modulated data symbol onto multiple sub-carriers also allows an increase in the symbol duration.
Since the throughput on each sub-carrier is greatly reduced, the symbol duration obtained through an OFDM scheme is
much larger than that of a single carrier modulation technique with a similar overall transmission bandwidth. In general,
when the channel delay spread exceeds the guard time, the energy contained in the ISI will be much smaller with
respect to the useful OFDM symbol energy, as long as the symbol duration is much larger than the channel delay
spread, that is:
T >> τ .
s
Although large OFDM symbol duration is desirable to combat time-dispersion caused ISI, however, the large OFDM
symbol duration can reduce the ability to combat the fast temporal fading, especially if the symbol period is large
compared to the channel coherence time. Thus, if the channel can no longer be considered as constant through the
OFDM symbol, the inter-sub-carrier orthogonality loss is introduced and the performance in fast fading conditions are
degraded. Hence, the symbol duration should be kept smaller than the minimum channel coherence time. Since the
channel coherence time is inversely proportional to the maximum Doppler shift f , the symbol duration T needs to be,
d s
in general, chosen such that:
T << .
s
f
d
4.1.6 Impact of Inter-Carrier Spacing
Because of the time-frequency duality, some of the time-domain arguments of clause 4.1.5 Impact of Symbol Duration
can be translated to the frequency domain in a straightforward manner. The large number of OFDM sub-carriers makes
the bandwidth of the individual sub-carriers small relative to the overall signal bandwidth. With an adequate number of
sub-carriers, the inter-carrier spacing is much narrower than the channel coherence bandwidth. Since the channel
coherence bandwidth is inversely proportional to the channel delay spread τ, the sub-carrier separation is generally
designed such that:
Δf << .
τ
In this case, the fading on each sub-carrier is frequency flat and can be modelled as a constant complex channel gain.
The individual reception of the QAM symbols transmitted on each sub-carrier is therefore simplified to the case of a
flat-fading channel. Moreover, in order to combat Doppler effects, the inter-carrier spacing should be much larger than
the maximum Doppler shift f :
d
Δf >> f .
d
4.1.7 OFDM Inactive Sub-Carriers
Since the OFDM sampling frequency is larger than the actual signal bandwidth, only a sub-set of sub-carriers is used to
carry QAM symbols. The remaining sub-carriers are left inactive prior to the IFFT, as illustrated in figure 6. The split
between the active and the inactive sub-carriers is determined based on the spectral constraints, such as the bandwidth
allocation and the spectral mask.
ETSI
13 ETSI TR 102 443 V1.1.1 (2008-08)

u
N
Inactive Inactive
Sub- Sub-
Carriers Carriers
N
Frequency (Sub-Carrier)
Figure 6: OFDM spectrum with inactive sub-carriers
The N modulated sub-carriers (i.e. carrying information), are centered in the N FFT bins, with the remaining inactive
u
sub-carriers, on either side of the modulated sub-carriers.
4.1.8 Time-Frequency Multiplexing
Multiple users can be multiplexed, both in time and in frequency, with pilot and signalling information. In the frequency
dimension (i.e. the sub-carrier dimension), users data symbol can be multiplexed on different numbers of useful
sub-carriers. In addition, sub-carriers or group of sub-carriers can be reserved to transmit pilot, signalling or other kind
of symbols. Multiplexing can also be performed in the time dimension, as long as it occurs at the OFDM symbol rate or
at a multiple of the symbol rate (i.e. from one IFFT computation to the other, every k*T seconds). The modulation
s
scheme (modulation level) used for each sub-carrier can also be changed at the corresponding rate, keeping the
computational simplicity of the FFT-based implementation. This allows 2-dimensional time-frequency multiplexing, of
the form shown in figure 7.
ETSI
Transmitted Power
14 ETSI TR 102 443 V1.1.1 (2008-08)

Frequency (Useful sub-carriers) →
D D D D D D D D D D D D D D D D D

4 4 4 4 4 4 4 4 4 6 6 6 6 6 6 6 6
D D D D D D D D D D D D D D D D D

4 4 4 4 4 4 4 4 4 6 6 6 6 6 6 6 6
D D D D D D D D D D D D D
P   P  P   P
4 4 4 4 4 4 4 6 6 6 6 6 6
D D D D D D D D D D D D D D D D D

4 4 4 4 4 4 4 4 4 6 6 6 6 6 6 6 6
D D D D D D D D D D D D D D D D D

4 4 4 4 4 4 4 4 4 6 6 6 6 6 6 6 6 ↓
D D D D D D D D D D D D D D D T

P  P
4 4 4 4 4 4 4 4 6 6 6 6 6 6 6 s
D D D D D D D D D D D D D D D D D ↑

4 4 4 4 4 4 4 4 4 6 6 6 6 6 6 6 6

D D D D D D D D D D D D D D D D D

2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2
D D D D D D D D D D D D D D D D D

2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2
D D D D D D D D D D D D D D D D D

2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2
D D D D D D D D D D D D
P   P D  P   P
2 2 2 2 2 2 2 2 2 2 2 2
D D D D D D D D D D D D D D D D D

2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2

P = pilot or signalling, D = data.
The subscript indicates the modulation level M=2, 4 or 6 (QPSK, 16QAM or 64QAM).

Figure 7: Example of OFDM 2-D structure
4.1.9 OFDM Signal Reception Using the FFT
At the receiver, a computationally efficient Fast Fourier Transform (FFT) is used to demodulate the multi-carrier
information and to recover the transmitted data.
4.2 OFDM for Mobile Terrestrial and Satellite Scenario
OFDM has intrinsic features that are generally acknowledged to be well suited to the terrestrial mobile radio
environment. In the case of S-DMB, these characteristics are useful in the Complementary Ground Component (CGC)
channel. In particular, the following characteristics are worth noting:
• Time dispersion: the use of several parallel sub-carriers in OFDM enables longer symbol duration, which
makes the signal inherently robust to time dispersion. Furthermore, a guard time may be added to combat
further the ISI.
• Spectral Efficiency: OFDM is constructed with fully orthogonal carriers, hence allowing tight frequency
separation and high spectral efficiency. The resulting spectrum also has good roll-off properties, given that
cross-symbol discontinuities can be handled through time windowing alone, filtering alone, or through a
combination of the two techniques.
ETSI
← Time
15 ETSI TR 102 443 V1.1.1 (2008-08)
• Reception: even in relatively large time dispersion scenarios, the reception of an OFDM signal requires only
an FFT implementation in the UE. No intra-cell interference cancellation scheme is required. Furthermore,
because of prefix insertion, OFDM is relatively insensitive to timing acquisition errors. On the other hand,
OFDM requires to perform frequency offset correction.
• Extension to MIMO: since the OFDM sub-carriers are constructed as parallel narrow band channels, the
fading process experienced by each sub-carrier is close to frequency flat and therefore, can be modelled as a
constant complex gain. This may simplify the implementation of a MIMO scheme if this is applied on a
sub-carrier or subset of carrier basis.
5 OFDM and the satellite environment
5.1 Non-Linearity Effects and Predistortion Techniques
When designing wireless communication systems and satellite links in particular, besides the impairments connected
with the presence of the radio channel, which can be both frequency and time selective causing strong linear distortion,
another severe source of degradation is introduced by High Power Amplifiers (HPA), which can cause non-linear
distortion in the transmitted signal, degrading the overall system performance. This occurs when the HPA is driven near
saturation, so as to exploit all the available output power and to increase power efficiency. This is particularly true for
the OFDM radio interface which is characterized by a rather high Peak-to-Average Power Ratio (PAPR). Besides these
factors, the cost of apparatus is another key issue: to properly exploit the expensive equipment, it is necessary to drive it
to the limit. This is certainly applicable to the satellite on-board HPA, but it is also true for ground terminals conceived
for mass-market, where slight cost reductions per device lead to large overall profits.
A consequence of these facts is that usually the impact of non-linear distortion on the transmitted signal is very strong,
as it acts directly on the band-limited pulse stream. The degradation includes amplitude and phase distortion, described
by AM/AM and AM/PM characteristics and the generation of in-band and out-band inter-modulation frequencies.
These phenomena lead to an increased Adjacent Channel Interference (ACI) due to a widening of the transmitted signal
spectrum. In particular, at the receiver each signal constellation point is warped and appears as a cluster, as ISI is
generated. These effects can be more or less penalizing for the system depending on the considered HPA characteristics
and on the distance from saturation.
The techniques able to counteract non-linear distortion are numerous and include the use of strong channel coding, the
use of equalization techniques at the receiver and the use of predistortion techniques at the transmitter. All of these
approaches try to mitigate the SNR loss for a given BER, allowing to increase the amplifier output power. Another
solution is obviously to back-off from saturation, but as seen before it is not desirable in the majority of cases, where
stringent power constraints exist.
The design of a NL compensator should consider a variety of factors, such as coding and modulation schemes, channel
estimator subsystem, system service requirements, DC-RF conversion efficiency constraints, system complexity and
cost, output power and adjacent channel interference specifications.
5.1.1 Compensation Techniques
In the scientific literature, several techniques have been proposed as means of mitigating non-linear distortion [i.4]:
• The simplest possibility is to back off from saturation, driving the HPA into a more linear region, at the
expense of a reduction of the available RF output power, making the link budget fulfilment difficult and of a
reduced DC-RF conversion efficiency. Clearly, this solution cannot be applied for on-board amplifiers, given
the stringent efficiency and link budget requirements. On the other hand, for on-ground gateway (GW)
amplifiers this is an easy way to avoid the unwanted non-linear effects.
• Another solution involves mitigation techniques at the receiver side. This can be efficiently achieved by using
equalizers, which try to compensate for the ISI and the constellation point warping. They represent a good
choice when there are strict complexity and cost constraints at the transmitter and complexity can be
concentrated at the receiver. The main drawback is that the signal is processed after the nonlinear distortion,
which hinders the possibility to eliminate the undesirable adjacent channel interference [i.5].
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16 ETSI TR 102 443 V1.1.1 (2008-08)
• In order to avoid the generation of adjacent channel interference, the compensation can be introduced before
the HPA, so that its output is a linearly amplified version of the original signal. This approach is commonly
referred to as predistortion, as it consists in processing the signal to be transmitted by means of a nonlinear
function, compensating the distortion introduced by the HPA [i.6].
5.1.2 Digital Predistortion Techniques
Predistortion techniques proposed in the literature can be divided into two main classes: digital predistorters and analog
predistorters. Essentially the waveform (analog) predistorter compensates for the memory less nonlinearity HPA and it
is placed after the pulse shaping filter in RF (or possibly IF) band; the digital predistorter is required to compensate a
nonlinearity with memory generated by the cascade of the linear pulse shaping filter, which introduces memory and the
HPA, which can be conceived as a memory less nonlinearity placed right after the pulse shaping filter at baseband. The
digital predistorter exhibits more flexibility in determining the predistorter coefficients than the analog counterpart,
since the learning algorithm is programmable in the digital predistorter. This suggests that the digital predistorter can be
more dynamically adaptive when system characteristics change as a consequence of a variation in the signal
characteristics (i.e. the pdf) or in the power amplifier characteristics (e.g. as a function of temperature, ageing or bias
fluctuations).
Digital predistortion techniques can be again subdivided into two categories, namely data constellation predistortion
and fractional (or oversampled) predistortion, according to the location of the compensator. Data constellation
predistorters are placed in the baseband system before the pulse shaping transmit filter, while fractional predistorters
are located after the pulse shaping transmit filter. Each of these predistortion techniques can be implemented so as to
accomplish a static compensation of HPA nonlinearity effects or to realize an adaptive compensation.
For this study, a fractional predistorter implemented by a gain-based LUT approach is considered [i.7]. The
compensator is located after the pulse shaping transmit filter and can correct the average positions of the individual
clusters and reduce their variance, bounding the effects of ISI. A square root raised cosine FIR filter is assumed as pulse
shaping filter. The output of the pulse shaping filter becomes the input of the predistorter, as in figure 8.
Gain based LUT fractional
b x y
predistorter
n m m
Pulse
HPA
shaping
Amplitude Phase
g
m
LUT
|.|
(F)
Figure 8: Gain-based LUT fractional predistorter block diagram
Linear in power LUT indexing will be considered, using table entries uniformly spaced along the input signal power
range, yielding denser table entries for larger amplitudes. It is characterized by a simple implementation, since it only
requires a square module computation and it is particularly effective if the non-linear effects are localized at large
amplitudes. The number on LUT entries depends on the allowed complexity at the transmitter, but typical values are
from 128 to 1 024. LUT entries computation is based on the inversion of the HPA characteristics that can be properly
modelled through analytic expressions.
5.1.3 Multi-Beam Coverage Using OFDM
One of the peculiar characteristics of OFDM, which is largely used in DVB-T, is the relative easiness in deploying
Single Frequency Networks (SFNs). This is achieved by synchronizing transmissions from various base stations and by
exploiting the guard time to resolve any residual asynchronicity in the signals received from different sources.
However, this works if and only if the guard time exceeds the relative delay difference between the two signals. This
implies in turn that the cell radius cannot be excessively large, to avoid very large guard times and hence overheads. In
other words, terrestrial SFNs based on OFDM have limited cell size and necessitate in general of a large number of base
stations.
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17 ETSI TR 102 443 V1.1.1 (2008-08)
It is very interesting to note that the application of OFDM in the forward link of a multi-beam antenna coverage from a
GEO or non GEO satellite can lead to a much simpler and more effective realization of a SFN. In fact, in a multi-beam
antenna coverage, interference from adjacent beams is generated from antenna sidelobes in the direction to the
interfered user. In essence, the desired signal and the interference follow exactly the same electro-magnetic path, except
for the on-board beamforming and antenna feed circuits. Therefore, the relative delays between desired signal and
interference are extremely small, if at all present. In conclusion, the guard time necessary for the realization of an SFN
through a multi-beam antenna coverage is much smaller than for the terrestrial case and the beam footprint size is not
limited in any way.
6 OFDM feasibility
6.1 Physical Layer Structure in the OFDM Downlink
6.1.1 Physical Channel
Physical channels are defined by a specific carrier frequency, set of orthogonal sub-carriers or sub-bands, time start &
stop (or duration), time-frequency interleaving pattern (possibly frequency hopping pattern). Given a carrier frequency,
physical channels are therefore mapped onto a specific 2-dimensional area in the time-frequency plane. Before time-
frequency interleaving, each physical channel corresponds to a set of sub-bands, while after symbol interleaving, the
sub-bands are distributed in a controlled manner across the overall frequency band. The time durations for specific time
units for the OFDM HS-DSCH are identical to those of 3GPP and can therefore be measured in integer multiples of
WCDMA chips, where the chip rate is 3,84 MHz. The time intervals defined in this configuration are:
• Radio frame: Also called an OFDM frame, a radio frame is a processing duration which consists of 15 slots.
The length of a radio frame corresponds to 38 400 chips (10 msec).
• Slot: A slot corresponds
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